Transmit circuit for imaging with ultrasound

ABSTRACT

A digital-to-analog converter with differential outputs is connected to two difference amplifiers through current splitters. The current splitters isolate the compliance voltage of the digital-to-analog converter so that larger resistances may be used with the difference amplifiers. The larger resistances allow for better signal-to-noise ratio performance of the transmit circuit. The difference amplifiers provide current signals to a push-pull output amplifier through their supply nodes. A single scaling resistor connects between the conventional outputs of two differential amplifiers to reduce mismatching between the positive and negative waveform paths. As a result of the feedback between the two difference amplifiers, a lower level of subharmonic and/or harmonic distortion products is achieved.

BACKGROUND

This invention relates to ultrasound transmit circuits. In particular,ultrasound transmit circuits for generating bi-polar ultrasoundwaveforms are provided.

For medical diagnostic ultrasound imaging, high current and high voltageamplifiers generate bi-polar waveforms. A wide bandwidth of operation ofthe amplifiers is used. However, operation of the transmit amplifiersmay generate even order distortion products, i.e., such a components ata second harmonic or subharmonics of the fundamental frequency of thebi-polar waveform. The subharmonics and/or harmonics generateundesirable echoes where an ultrasound system is designed to receivevaluable information generated by tissue.

Push-pull amplifiers have been used to reduce even order distortion intransmitted ultrasound waveforms. FIG. 1 shows a push-pull amplifier 100disclosed in U.S. Pat. No. 3,895,306. The push-pull amplifier 100includes two class A cascode amplifiers 102 and 104 connected in apush-pull relationship with an output transformer 106. Each cascodeamplifier includes two transistors 108, 112 and 110, 114. Twotransistors 108, 110 are connected in a common base configuration, andthe other two transistors 112, 114 operate as common emitter stages.

A network 116 of resistors 124, 126, 138 and a capacitor 118 provides afeedback loop for the push-pull amplifier 100. The network 116 detectsdifferences in the output of the two cascode amplifiers 102, 104 as afunction of the current at the center tap 132 of the transformer 106.Any difference at the center tap 132 generates a voltage at the bases ofthe transistors 108, 110. The voltage is applied to the bases of thetransistors 112 and 114 through the capacitor 118 to equalize thecascode amplifiers transfer function. The network 116 provides negativefeedback, and the resistors 124, 126 and 138 establish a DC operatingvoltage. Two input sources 128, 130 provide a signal of the sameamplitude but 180° out of phase to the cascode amplifiers 102, 104. Aninductor 134 isolates the feedback path for the network 116 from asupply voltage 136.

Harmonic and/or subharmonic distortion produced by the two cascodeamplifiers 102 and 104 are substantially identical when the fundamentaloutput of the cascode amplifiers 102, 104 are of the same amplitude. Thesignals output by the cascode amplifiers 102, 104 are equal andopposite. Any even harmonics generated by the cascode amplifiers 102,104 are cancelled in the output transformer 106. However, any differencein the fundamental waveforms generates a feedback signal. The feedbacksignal is in phase with respect to the branch with the lower outputamplitude and out of phase for the branch with the higher outputamplitude. The feedback signal tends to equalize the output of the twobranches of the push-pull amplifier 100.

This push-pull amplifier 100 is a Class A amplifier. Class A amplifiershave high quiescent power dissipation, resulting in low efficiency.Higher efficiency is achieved by Class B amplification. For Class Bamplification, each path provides output for alternate time periods. Thepositive and negative portions of the bi-polar waveform are separatedfor amplification. Consequently, subharmonic and/or harmonic distortionsin a Class B amplifier cannot be cancelled by the feedback signal. Toreduce these distortions, the two paths are matched in gain and phase.

A high efficiency linear transmit circuit for ultrasound diagnosticimaging is disclosed in U.S. Pat. No. 6,104,673 and is shown in FIG. 2.The transmit circuit 200 operates over a wide frequency bandwidth. Thetransmit circuit 200 includes a programmable waveform generator (PWG)202, two digital-to-analog converters 210, 212, a respective pair ofcurrent amplifiers or drivers 214, 216 and an output amplifier 218. Theoutput amplifier includes a pair of transistors 222 and 224, and atransformer 220.

The PWG 202 generates separate unipolar waveforms representing positiveand negative portions of the desired bi-polar ultrasound waveform. Oneunipolar waveform is output on bus 206 to a digital-to-analog converter212, and the other unipolar waveform is output on bus 208 todigital-to-analog converter 210. A sign bit is output on line 204 toenable operation of the digital-to-analog converters 210 and 212. Thetwo transistors 222 and 224 are connected in a common gateconfiguration. An external voltage source 226 provides gate biasing. Acenter tap of the primary winding of the transformer 220 is tied to ahigh voltage power supply 228. Since the transmit circuit 200 includestwo open loop signal paths for respective positive and negative portionsof the bi-polar transmit waveform, the components in each path should beclosely matched to avoid harmonic and/or subharmonic distortion.

In order to transmit a waveform with a Gaussian envelope (FIG. 4A), thecurrent-output DACs 210 and 212 are intended to produce a pair ofsignals shown in FIGS. 4B and 4C, respectively. Having ideally matchedsignal paths, transmit signal, U(t), is combined as the algebraicdifference of positive, U⁺(t), and negative, U⁻(t), portions inaccordance with:

U(t)=U ⁺(t)−U ⁻(t)  (1)

Assume further that there is a gain mismatch between the two signalpaths, denoted as δ=ΔG/G. In such a case, a “distorted” transmit signal,U_(D)(t), yields

U _(D)(t)=U(t)+δ[U ⁺(t)+U ⁻(t)]  (2)

The second term of Equation 2 will produce even order distortionproducts. For instance, given the waveform with the Gaussian envelope,the resulting spectrum expands as shown in FIG. 4D.

In practice, the purity of a transmitted waveform is estimated with theLinear Response Rejection Ratio (LRRR). The LRRR is defined as the ratioof the energy under matched filters that are centered at fundamental andthe second harmonic frequencies. For a Gaussian envelope, the LRRR canbe easily computed. The obtained results (FIG. 4E) show that the priorart transmit cell 200 is quite sensitive to the gain mismatch. Using adual DAC topology has a significant drawback since the level of gainmismatch is twice as much higher. This is particularly meaningfulbecause DACs, even high-resolution DACs, may have gain error up to few %of the full scale.

BRIEF SUMMARY

The present invention is defined by the following claims, and nothing inthis section should be taken as a limitation on those claims. By way ofintroduction, the preferred embodiments described below include a methodand system for generating a bi-polar ultrasound transmit waveform. Adigital-to-analog converter with differential outputs is connected totwo difference amplifiers by current splitters. The differenceamplifiers provide current signals to the push-pull output amplifier forgenerating a desired bi-polar ultrasound waveform. A resistor connectingbetween the conventional outputs of two differential amplifiersspecifies the voltage-to-current scaling factor for both amplifiers.Employing a single resistor, both positive and negative portions of awaveform are uniformly processed. The current splitters allow thedigital-to-analog converter to have a low compliance voltage, such as0.2 or 0.3 volts for an integrated converter, while the differenceamplifiers operate at higher voltages for better signal-to-noise ratioperformance.

In a first aspect, an ultrasound transmit circuit for generating abi-polar waveform is provided. An output of a digital-to-analogconverter connects with a first current splitter. An ultrasoundtransducer operatively connects to receive a signal responsive to thedigital-to-analog converter.

In a second aspect, an ultrasound transmit circuit for generating abi-polar waveform includes an output amplifier having first and secondinputs. First and second difference amplifiers have respective first andsecond supply nodes connected with the first and second inputs,respectively. An ultrasound transducer connects with the outputamplifier. First and second current splitters connect with inputs of thefirst and second difference amplifiers. A digital-to-analog converterhas differential outputs. A first one of the differential outputsconnects with the first current splitter, and a second one of thedifferential outputs connects with the second current splitter.

In a third aspect, a method for generating a bi-polar ultrasoundwaveform with an ultrasound transmit circuit is provided. A current froma digital to analog converter is split. An output amplifier is driven inresponse to at least part of the split current. The bi-polar ultrasoundwaveform in response to the driven output amplifier.

Further aspects and advantages of the invention are discussed below inconjunction with the preferred embodiments.

BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS

FIG. 1 is a block diagram of a Class A amplifier used for an ultrasoundtransmit circuit.

FIG. 2 is a circuit diagram of a Class B amplifier used for anultrasound transmit circuit.

FIG. 3 is a circuit diagram of one embodiment of an ultrasound transmitcircuit.

FIG. 4A shows one embodiment of a transmit waveform.

FIGS. 4B and 4C are driving signals for a push-pull Class B stage toproduce the waveform shown in FIG. 4A.

FIG. 4D illustrates distortion of a transmit spectrum caused by the gainmismatch using the waveforms of FIGS. 4A-C in the circuit of FIG. 2.

FIG. 4E is a graph of Linear Response Rejection Ratio verses gainmismatch using the waveforms of FIGS. 4A-C in the circuit of FIG. 2.

FIG. 5 is a circuit diagram of another embodiment of an ultrasoundtransmit circuit.

FIG. 6 is a circuit diagram of yet another embodiment of an ultrasoundtransmit circuit.

FIG. 7 is a graphical representation of the effects of a mismatch.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Several embodiments provide reduced level of harmonic and/or subharmonicdistortion for class B ultrasound amplifiers. In a first embodiment, asingle digital-to-analog converter is used to avoid inaccuraciesassociated with different full-scale amplitudes caused by separatedigital-to-analog converters for each path. A switch connects the outputof the digital-to-analog converter to one of two current drivers fordriving a Class B amplifier. The current drivers include amplifiers withfeedback, allowing control of current gain in the current driver.

In a second embodiment, a single digital-to-analog converter with adifferential output connects with two difference amplifiers. Supplynodes of the differential amplifiers drive a Class B output amplifier. Aresistor connects between the conventional outputs of the twodifferential amplifiers for uniform scaling the two paths, avoidinggeneration of harmonic and/or subharmonic distortion. To allow fordigital-to-analog converters with lower compliance voltages, currentsplitters connect between each of the outputs of the digital-to-analogyconverters and the differential amplifiers.

Both embodiments provide broadband high power ultrasound transmitamplifiers effective for reducing even order distortion. Arbitrarywaveforms may be amplified by this low cost circuitry. Initial balancingof circuits is avoided. The circuit topology minimizes gain and phasemismatch between two signal paths driving the push-pull outputamplifier. Different load impedances may also be connected withoutgenerating harmonic and/or subharmonic distortion.

FIG. 3 shows an ultrasound transmit circuit 500 using a switch 510. Inone embodiment, the ultrasound transmit circuit 500 is used for medicaldiagnostic ultrasound imaging. The ultrasound transmit circuit 500includes a programmable waveform generator (PWG) 502, adigital-to-analog converter (DAC) 508, the switch 510, first and secondcurrent drivers 512, 514, an output amplifier 516 and a transducerelement 517. Additional, fewer or different components may be used.

The PWG 502 comprises a memory and associated circuitry for generating adigital representation of an ultrasound transmit waveform. In oneembodiment, the PWG 502 comprises the circuitry shown in U.S. Pat. No.6,104,673, the disclosure of which is incorporated herein by reference.For example, a memory, multipliers, delay buffers, adders, and othercircuitry generate a digital representation of a modulated bi-polarultrasound waveform from envelope samples output from the memory. Inalternative embodiments, a processor, digital signal processor,application specific integrated circuit, or memory output any subset orentire representation of the bi-polar ultrasound waveform to the DAC508.

The PWG 502 outputs the digital representation to the DAC 508 on a bus506. The PWG 502 also outputs a switch control signal on the line 504 tothe switch 510. The switch control signal comprises a polarity sign bit.The polarity sign bit distinguishes between positive and negativeportions of the digital representation of the bi-polar waveform. Forexample, the PWG 502 generates a digital representation comprising aunipolar waveform where portions of the unipolar waveforms represent themagnitude (module) of negative and positive portions of the desiredbi-polar ultrasound waveform.

The DAC 508 comprises a current output single ended digital-to-analogconverter. Other digital-to-analog converters may be used. In oneembodiment, the DAC 508 is operable to receive 8 bits of datarepresenting an amplitude at any given time, but different resolutionsmay be provided. The data on the output bus 506 is converted into ananalog ultrasound waveform by the DAC 508.

The switch 510 comprises a high-speed single-pole double-throw analogswitch (FET or CMOS), 2-channel multiplexer, a differential amplifier,or other switch device. In this embodiment, the switch includes an input518 and two outputs 520 and 522. The switch 510 receives the analogwaveform from the digital-to-analog converter 508.

The switch 510 also includes a control signal input 524. The polarityinformation controls the switch 510. In response to the polarity signal,the switch 510 selects between the two outputs 520 and 522. The portionof the waveform representing the positive portion of the desiredbi-polar transmit waveform is routed to the first current driver 512,and the portion of the waveform representing the negative portion of thedesired bi-polar waveform is routed to the second current driver 514.

The first and second current drivers 512 and 514 receive signalscorresponding to positive and negative portions respectively, of thedesired bi-polar waveform. Each current driver 512, 514 comprises aclosed loop current mirror. In one embodiment, each of the currentdrivers 512, 514 comprise an operational amplifier 526, 528, tworesistors 534, 536, 548 and 550 and a transistor 530, 532. Thetransistors 530, 532 act as series pass devices. One resistor 548, 550connected between the switch 510 and ground comprises a load resistor,and another resistor connected between the operational amplifier andground in a feedback loop comprises a scaling resistor. The gain of thecurrent driver is a function of the ratio of the load resistor to thescaling resistor. In one embodiment, the scaling resistor has a lowresistance, such as 0.5 to 2 ohms, for use with high current, such as afew amps. Other currents and resistances may be used. In one embodiment,unity gain or a gain of 1 is used, but other gains may be provided. Theresistors in each of the current drivers 512, 514 are matched to theresistances of the other current driver 512, 514.

The transistors 530, 532 comprise a MOSFET transistor connected in acommon gate configuration. In an alternative embodiment, a bipolartransistor in a common base configurations is used. In other alternativeembodiments, different current drivers, such as conventional currentmirrors, are provided.

Each current driver 512, 514 comprises a path or channel for driving theoutput amplifier 516. One path is responsive to waveforms representingthe positive portion of the desired bi-polar ultrasound waveform, andthe other path is responsive to waveforms representing the negativeportion of the desired bi-polar ultrasound waveform. In one embodiment,the input waveforms representing both the positive and negative portionsof the bi-polar waveform are unipolar positive waveforms. Alternatively,unipolar negative waveforms may be input.

The output amplifier 516 comprises a Class B push-pull amplifier stageThe output amplifier 516 is a transformer-coupled amplifier, including atransformer 538, two transistors 540, 542 and two voltage supplies 546and 544. The transformer 538 may have any winding ratio, such as a 1.5to 1 step down ratio. The center tap of the primary winding of thetransformer 538 connects with a DC voltage source 546. Since theunipolar waveforms representing respective positive and negativeportions of the bi-polar waveform are applied to different sides of thetransformer 538, the bi-polar ultrasound waveform is generated at theoutput of the transformer 538.

The voltage source 546 connected to the transformer 538 comprises a highvoltage DC power supply. For example, the voltage source 546 comprises avoltage divider, a battery, or other source of DC voltage. In oneembodiment, the voltage source 546 provides about 100 volts, but otherhigher or lower voltages may be provided.

The transistors 540, 542 comprise MOSFETS or bi-polar junctiontransistors. Other types of transistors may be used. The transistors 540and 542 are connected in a common gate (base) configuration. Thetransistors 540 and 542 are biased by the voltage source 544.

The voltage source 544 connected to the transistors 540, 542 comprises avoltage divider, a battery, or other device for providing biasing DCvoltage. In one embodiment, the bias voltage is 10 to 15 volts, buthigher or lower biasing voltages may be used.

The transistor 540, 542 of the output amplifier 516 and the transistors530, 532 of the current drivers 512, 514 comprise two cascodeamplifiers. In effect, a cascode amplifier is provided in each of twochannels associated with respective ones of the current drivers 512,514. The cascaded transistors 530, 532, 540, 542 provide a lowcapacitive load to the operational amplifiers 526, 528, expanding theeffective bandwidth of the operational amplifiers 526, 528 or allowingamplifiers with lesser driving capabilities.

The transducer element 517 receives the bi-polar ultrasound waveformfrom the output amplifier 516. The transducer element 517 comprises apiezoelectric, micro-electromechanical membrane, or other transducerdevice for converting electrical energy to acoustic energy. In oneembodiment, the transducer element 517 is one of an array of transducerelements. Each element of the array is connected with a different outputamplifier.

In operation, the transmit circuit 500 generates a bi-polar ultrasoundwaveform, such as shown in FIG. 4A. The unipolar waveforms shown inFIGS. 4B and 4C are combined in the PWG 502 and their superpositionprovided by the digital-to-analog converter 508. In alternativeembodiments, negative unipolar waveforms are generated. The unipolarwaveform shown in FIG. 4B represents the positive portions of thedesired bi-polar waveform, and the unipolar waveform shown in FIG. 4Crepresents the negative portions of the desired bi-polar waveform shownin FIG. 4A. Arbitrary waveforms having different envelopes, amplitudesor frequencies may be used.

The PWG 502 generates data corresponding to a unipolar waveformrepresenting the desired bi-polar ultrasound waveform. Bits of datarepresenting magnitude information are provided to the digital-to-analogconverter 508. The digital-to-analog converter 508 applies an equivalentanalog current to the input terminal 518 of the switch 510.

The switch 510 also receives a sign control bit or indication of thepositive or negative portion of the bi-polar waveform represented by theunipolar waveform. In response to this indication, the switch 510switches between the two outputs 520, 522. For example, the unipolarwaveform of FIG. 4B is switched to one current driver 512. The otherportion of the unipolar waveform as represented in FIG. 4C is switchedto the other current driver 514.

One current driver 512 amplifies the unipolar waveform representing thepositive portion of the desired bi-polar waveform, and the other currentdriver 514 amplifies the unipolar waveform representing the negativeportion of the desired bi-polar waveform. When not amplifying thecurrent, each current driver 512, 514 is grounded through the loadingresistor 548, 550. Both amplified currents are provided to the outputamplifier 516.

The output amplifier 516 transforms the two unipolar waveforms into thebi-polar ultrasound waveform shown in FIG. 4A or another bipolarwaveform. The unipolar waveform representing the positive portion of thebi-polar waveform is provided to one end of the primary winding of thetransformer 538, and the unipolar waveform representing the negativeportion of the desired bi-polar waveform is provided to the other end ofthe primary winding of the transformer 538. Being fed by the voltagesource 546 at the center tap of the primary winding, the transformer 538outputs the desired bi-polar ultrasound waveform.

The closed loop current mirrors or current drivers 512, 514 providestable and properly matched current gain for both signal paths as afunction of the scaling resistors 534, 536. Well matched current gainreduces even order harmonic distortions. By calibrating the resistancesof 534, 536, the difference between two paths may be additionallycorrected.

To further reduce even order distortion, a topology with a singlescaling resistor is provided. One embodiment using a single scalingresistor and a single DAC is shown in FIG. 5. The ultrasound transmitcircuit 600 uses similar components as the ultrasound transmit circuit500 of FIG. 3. The discussion of these common components for FIG. 3applies to this discussion for FIG. 5. The transmit circuit 600 includesa programmable waveform generator 602, a digital-to-analog converter604, current drivers 608, 610, a scaling resistor 612, an outputamplifier 606, and a transducer element 607. Additional, fewer ordifferent components may be used.

The PWG 602 comprises the circuitry disclosed in U.S. Pat. No. 6,104,673altered to output digital data representing a bi-polar transmit signal,arranged by an offset binary code. In the analog domain, the bi-polartransmit signal corresponds to an unipolar waveform with a DC offset.Other coding or data representing bipolar signals may be provided.

The DAC 604 comprises a communications DAC or other digital-to-analogconverter with two complementary outputs. The complementary outputscomprise differential outputs of the inverse of the same waveform. Inalternative embodiments, a digital-to-analog converter 604 with a singleoutput and an inverter connected with the output may be used to providethe differential outputs.

Each of the current drivers 608, 610 comprises a difference amplifier.Each difference amplifier comprises an operational amplifier 624, 626and two pairs of match resistors labeled R1 and R2. In one embodiment,the resistors R1 are about 100 ohms, but other resistance values may beused. The resistors R2 are 100 to 1K ohms, but other resistance valuesmay be used. A unity gain with respect to both the inverted andnon-inverted inputs from the digital-to-analog converter 604 isprovided. The difference amplifiers comprise level shift circuitsconverting the differential signal provided by the digital-to-analogconverter 604 to a single ended bipolar output relative to the ground.

The inverted output of the digital-to-analog converter 604 is providedas a positive input to one of the operational amplifiers 626 and as anegative input to the other operational amplifier 624. The non-invertedoutput of the digital-to-analog converter 604 connects to the otherinputs of the two operational amplifiers 624, 626. The current driversinclude inverted input 628, 630 and non-inverted inputs 632, 634corresponding to the connections with the operational amplifiersdiscussed above. With the unity gain, the output 636, 638 of each of theoperational amplifiers 624, 626 is zero if both input voltages areequal. For the offset binary code output by the PWG 602, thedifferential outputs of the DAC 604 have a same voltage or current whenthe transmitted waveform crosses zero. Given the desired bi-polarwaveform, the conventional outputs 636, 638 of the difference amplifiersof the current drivers 608, 610 comprise bi-polar waveforms centeredaround 0. Since the current driver 608 and 610 are driven in reverse,their conventional outputs 636, 638 are in opposite polarity.

The scaling resistor 612 connects between the conventional outputs 636and 638 of the operational amplifiers 624, 626. In one embodiment, aresistor with 1 to 5 ohms of resistance is directly connected betweenthe outputs 636, 638. In alternative embodiments, different resistancesor additional network elements may be connected with the scalingresistor 612. The single scaling resistor 612 or network provide a sameor substantially same transfer function for each of the current driver608, 610.

The operational amplifiers 624 and 626 include positive supply nodes 640and 642 and negative supply nodes 644 and 646. The negative supply node644 and 646 are connected to a common voltage source or differentvoltage sources providing a same or substantially same voltage. Thepositive supply nodes 640, 642 are connected to drive the outputamplifier 606. In alternative embodiments, the negative supply nodes644, 646 connect with the output amplifier 606, and the positive supplynode 640, 642 connect with a common voltage source.

The positive supply nodes 640, 642 comprise bias ports that provideoppositely phased current signals to drive the output amplifier 606. Theoperational amplifiers 624, 626 separate their output current intopositive and negative components using complementary Class B outputstages. Positive and negative components of the input waveform areseparated onto respective positive and negative supply nodes 640, 642,644, 646. The magnitude of the current is responsive to the resistanceof the scaling resistor 612. Since the output 636, 638 of theoperational amplifier 624 and 626 are in reverse or opposite phase ofeach other, the positive supply nodes 640 and 642 drive a pair ofin-phase current signals associated with positive and negative portionsof the desired bi-polar ultrasound waveform.

The output amplifier 606 receives the driving current signals that areequalized for gain and phase in response to the two substantiallyidentical closed loop difference amplifiers of the current driver 608and 610. In response, the output amplifier 606 generates the desiredbi-polar ultrasound waveform. The ultrasound waveform is provided to thetransducer element 607 for transmission of acoustic energy.

The scaling resistor 612 scales the current provided to the outputamplifier 606 and provides more efficient matching of the gain of thetwo current driver 608 and 610. Using a single digital-to-analogconverter 604 and the scaling resistor 612 connected with both currentdrivers 608 and 610, even order distortion generated by the transmitcircuit 600 is reduced.

Various components, such as the active components of the current drivers608 and 610, output amplifier 606 and/or DAC 604, are integrated in oneembodiment. For example, an application specific integrated circuitincludes the various active components.

The compliance voltage for an integrated DAC 604 or a separate DAC 604limits the input signal range applied to the current drivers 608 and610. For example, the compliance voltage may be rated at 1 volt orbelow, such as a 0.2 or 0.3 volt rating for a integrated current sourceDAC operable at high frequencies. Setting the scaling resistor 612appropriately, the above limitation may not adversely effect thetransmitter output span. A large span of output voltages is desired forultrasound imaging where the transmitter circuit 600 is used for bothhigh voltage B-mode imaging and lower voltage continuous wave (CW)Doppler imaging. To provide larger voltage outputs, the resistances R1and R2 are increased. However, the compliance voltage limits theresistances that may be used. As the compliance voltage becomes lower,the current driver's spot noise results in deterioration of the outputsignal-to-noise ratio (SNR). The SNR deterioration has less adverseeffects for high voltage B-mode transmissions than CW Dopplertransmissions. CW Doppler normally operates at lesser power levels(e.g., up to −20 . . . 30 dB in voltage). In such a case, the SNRreduction due to the inadequate input signal range may be noticeable.

FIG. 6 shows a transmit circuit 700 similar to the transmit circuit 600of FIG. 5, but with signal splitters 648, 650 between the differentialoutputs of the DAC 604 and the drivers 608, 610. This embodiment of thetransmit circuit 700 uses the single scaling resistor 612 and the singledifferential DAC 604 as shown in FIG. 5. The current drivers 608 and 610of the transmit circuit 700 connect with the output amplifier 606 asdiscussed above for the transmit circuit 600 of FIG. 5. For brevity, thedescription of common components, their alternatives andinterconnections is not repeated. The discussion below addresses thedifferences between the transmit circuit 700 of FIG. 6 and the transmitcircuit 600 of FIG. 5.

Each of the current splitters 648 and 650 are a pair of transistors 652.Any of various now know or later developed transistors 652, such as PNP,NPN, Mosfet, bi-polar or other transistors may be used. In oneembodiment, each current splitter 648 and 650 is a pair of matchedtransistors 652, but unmatched transistors 652 may be used. The pair oftransistors 652 of each current splitter 648, 650 have a common emitterterminal, a common base terminal, and two collector terminals. The pairof transistors 652 have commonly connected emitters and bases. Thecommon base terminals are grounded in a grounded-base configuration. Thecommon emitter terminals connect to one of the outputs of the DAC 604. Apositive input 632 of the first operational amplifier 624 connects withthe collector terminal of the one current splitter 648 while thenegative input 628 connects with the collector terminal of the othercurrent splitter 650. The second operational amplifier 626 employs thesame type of coupling as shown.

In operation, the DAC 604 in combination with the current splitters 648and 650 act as cascode amplifiers. The DAC 604 drives the emitterterminals while the base terminals are grounded. The transistors 652 ofthe current splitters 648 are always on. The DAC current is splitbetween the two collector terminals of the corresponding currentsplitter 648, 650.

Each of the current drivers 608, 610 is an operational amplifier 624,626 and a pair of matched resistors labeled R. The resistors R are 100to 1K ohms. The resistors R1 and R2 of FIG. 5 between the DAC 604 andthe operational amplifiers 624 and 626 are not provided, but areincluded in alternative embodiments. The operational amplifiers 624 and626 include positive supply nodes 640 and 642 and negative supply nodes644 and 646. The positive supply nodes 640, 642 are connected to drivethe output amplifier 606. The negative supply nodes 644 and 646 connectto a common voltage source V₃. The resistors R connected with thepositive input 632 and 634 connect to a biasing voltage source V₄. Theabsolute value of V₄ is less than the absolute value of V₃. For example,V₄ is −3 volts and V₃ is −5 volts. In alternative embodiments, V₄ is azero or ground potential.

The inputs of the operational amplifiers 624 and 626 are cross-coupled.The inverted output of the DAC 604 is provided as a positive input toone of the operational amplifiers 626 and as a negative input to theother operational amplifier 624. The non-inverted output of the DAC 604connects to the other inputs of the two operational amplifiers 624, 626.Both the inverted and non-inverted inputs of the operational amplifiers624, 626 convert the input current to an output voltage, Thedifferential current signal provided by the DAC 604 is translated tosingle-ended bipolar signals at terminals 636 and 638 relative to thebiasing power supply V₄.

The current splitters 648 and 650 allow large resistances R for thecurrent drivers 608, 610 without SNR problems due to the compliancevoltage limitations of the DAC 604. Driving a common-base stage, the DACoutput voltage is kept within a few tenths of a volt. Given the DACcurrent range, the voltage span at the collectors of the currentsplitters 648 and 650 is directly proportional to the resistance R.Increasing R improves the resulting SNR.

An ideally matched transistor pair 652 equally splits the current fromthe DAC 604 (e.g., a 50/50 division of the DAC's output current).Alternatively, the transistors 652 in one or both of the currentsplitters 648 and 650 mismatch. Any mismatch causes common-modelimitations, so no or little current in the resistor 612 connecting theoutputs of the operational amplifiers is generated.

Let I_(M) denote the full-scale output range of the differential DAC 604for each output. The provided differential signal crosses zero at anI_(M)/2 point. In such a case, the inverted and non-inverted outputs ofthe DAC 604, I₁ and I₂, are expressed as: $\begin{matrix}\begin{matrix}{I_{1} = {\frac{I_{M}}{2} - \delta}} \\{I_{2} = {\frac{i_{M}}{2} + \delta}}\end{matrix} & (1)\end{matrix}$

where δ is the magnitude of an alternating current component referred toa DC bias of I_(M)/2. In operation, δ≦I_(M)/2. Thus, if δ=I_(M)/2→I₁=0and I₂=I_(M).

For a 50/50 split, the inverted and non-inverted inputs of bothdifference amplifiers 608 and 610 are driven by I₁/2 and I₂/2 ininverse. In practice, a spitting error, γ, is caused by a mismatch. FIG.7 shows the currents where the splitting error range is limited by|γ|_(MAX)=½. Substituting Equation 1, a nodal analysis at the invertedand non-inverted inputs yields: $\begin{matrix}{\begin{matrix}{U_{1} = {\left\lbrack {{\left( {\frac{I_{M}}{2} - \delta} \right) \cdot \left( {\frac{1}{2} + \gamma_{2}} \right)} - {\left( {\frac{I_{M}}{2} + \delta} \right) \cdot \left( {\frac{1}{2} - \gamma_{1}} \right)}} \right\rbrack \cdot R}} \\{U_{2} = {\left\lbrack {{\left( {\frac{I_{M}}{2} + \delta} \right) \cdot \left( {\frac{1}{2} + \gamma_{1}} \right)} - {\left( {\frac{I_{M}}{2} - \delta} \right) \cdot \left( {\frac{1}{2} - \gamma_{2}} \right)}} \right\rbrack \cdot R}}\end{matrix}{{or}\text{:}}} & (2) \\\begin{matrix}{U_{1} = {\left\lbrack {{- \delta} + {\frac{I_{M}}{2} \cdot \left( {\gamma_{1} + \gamma_{2}} \right)} + {\delta \cdot \left( {\gamma_{1} - \gamma_{2}} \right)}} \right\rbrack \cdot R}} \\{U_{2} = {\left\lbrack {{+ \delta} + {\frac{I_{M}}{2} \cdot \left( {\gamma_{1} + \gamma_{2}} \right)} + {\delta \cdot \left( {\gamma_{1} - \gamma_{2}} \right)}} \right\rbrack \cdot R}}\end{matrix} & (3)\end{matrix}$

Equation 3 shows that the splitting errors cause a direct current offsetat the output nodes 636, 638 of the operational amplifiers 624 and 626.The amount of this offset, U_(OFF), is $\begin{matrix}{U_{OFF} = {\left\lbrack {{\frac{I_{M}}{2} \cdot \left( {\gamma_{1} + \gamma_{2}} \right)} + {\delta \cdot \left( {\gamma_{1} - \gamma_{2}} \right)}} \right\rbrack \cdot R}} & (4)\end{matrix}$

The transmit current is defined by the voltage across the scalingresistor R_(S). This voltage is:

U ₁ −U ₂=−2·R·δ  (5)

The transmit circuit 700 has zero or little sensitivity to splittingerrors. To avoid clipping the output, the supply voltages V₂ and V₃ arerated to provide room for any expected DC offset.

While the invention has been described above by reference to variousembodiments, it would be understood that many changes and modificationscan be made without departing from the scope of the invention. Forexample, different current drivers or output amplifier configurationsmay be used. Different digital waveforms generators may be used forproviding a digital representation to the digital-to-analog converter.Additional amplifiers and transistors may also be used. The polarity ofthe supply nodes of the differential amplifiers and associated voltagesources may be reversed. Likewise, the polarity of the cross-coupledinputs of the differential amplifiers or the outputs of the DAC may bereversed. Any of various analog and/or digital devices may be added,such as connecting between components described above.

It is therefore intended that the foregoing detailed description beunderstood as an illustration of the presently preferred embodiments ofthe invention, and not as a definition of the invention. It is only thefollowing claims or added claims, including all equivalents, that areintended to define the scope of this invention.

What is claimed is:
 1. An ultrasound transmit circuit for generating abi-polar waveform, the circuit comprising: a first current splitter; adigital-to-analog converter having an output connected with the firstcurrent splitter; and an ultrasound transducer operatively connected toreceive a signal responsive to the digital-to-analog converter.
 2. Thecircuit of claim 1 further comprising a second current splitter whereinthe digital-to-analog converter comprises a differentialdigital-to-analog converter, the first current splitter connected withone of a plurality of differential outputs of the digital-to-analogconverter and the second current splitter connected with another of theplurality of differential outputs of the digital-to-analog converter. 3.The circuit of claim 2 wherein a first positive input of a firstoperational amplifier connects with a first collector terminal of thefirst current splitter while a first negative input of the firstoperational amplifier connects with a first collector terminal of thesecond current splitter.
 4. The circuit of claim 3 wherein a secondpositive input of a second operational amplifier connects with a secondcollector terminal of the second current splitter while a secondnegative input of the second operational amplifier connects with asecond collector terminal of the first current splitter.
 5. The circuitof claim 1 wherein the first current splitter comprises first and secondtransistors with commonly connected emitters and bases where thecommonly connected bases are grounded and where the commonly connectedemitters connect to the output of the digital-to-analog converteroutput.
 6. The circuit of claim 1 further comprising an output amplifierconnected between the first current splitter and the ultrasoundtransducer.
 7. The circuit of claim 1 further comprising first andsecond current drivers connected between the first current splitter andthe ultrasound transducer.
 8. The circuit of claim 7 further comprisinga resistor connected between the first and second current drivers. 9.The circuit of claim 8 further comprising a transformer-coupledamplifier connected with the first and second current drivers; whereinthe first and second current drivers comprise first and seconddifference amplifiers, respectively, where a positive input of the firstdifference amplifier connects with the first current splitter and anegative input of the second difference amplifier connects with thefirst current splitter; wherein the resistor connects with conventionaloutputs of the first and second difference amplifiers, respectively; andwherein first and second supply nodes of the first and second differenceamplifiers, respectively, connect with the transformer-coupledamplifier, and third and fourth supply nodes of the first and seconddifference amplifiers, respectively, connect with a voltage source. 10.The circuit of claim 8 wherein a positive input of the first currentdriver connects to the first current splitter and to a resistor; andfurther comprising a voltage source connected with the resistor.
 11. Anultrasound transmit circuit for generating a bi-polar waveform, thecircuit comprising: an output amplifier having first and second inputs;first and second difference amplifiers having respective first andsecond supply nodes connected with the first and second inputs,respectively; an ultrasound transducer connected with the outputamplifier; first and second current splitters connected with inputs ofthe first and second difference amplifiers; and a digital-to-analogconverter having differential outputs, a first one of the differentialoutputs connected with the first current splitter and a second one ofthe differential outputs connected with the second current splitter. 12.The circuit of claim 11 wherein the first current splitter comprisesfirst and second transistors and the second current splitter comprisesthird and fourth transistors, the first one of the differential outputsconnected with the first and second transistors, the second one of thedifferential outputs connected with the third and fourth transistors,the first transistor connected with a positive input of the firstdifference amplifier, the second transistor connected with a negativeinput of the second difference amplifier, the third transistor connectedwith a negative input of the first difference amplifier, and the fourthtransistor connected with a positive input of the second differenceamplifier.
 13. The circuit of claim 11 further comprising a resistorconnected between first and second outputs of the first and seconddifference amplifiers, respectively.
 14. The circuit of claim 11 furthercomprising a first resistor connected between an input of the firstdifference amplifier and a voltage source and a second resistorconnected between an input of the second difference amplifier and thevoltage source.
 15. A method for generating a bi-polar ultrasoundwaveform with an ultrasound transmit circuit, the method comprising: (a)splitting a current from a digital to analog converter; (b) driving anoutput amplifier in response to at least part of the split current; and(c) generating the bi-polar ultrasound waveform in response to (b). 16.The method of claim 15 wherein (a) comprises splitting the current atabout a 50/50 ratio.
 17. The method of claim 15 wherein (b) comprisesdriving the output amplifier with supply currents of first and seconddifference amplifiers responsive to the split current; and furthercomprising (d) scaling supply currents of the first and the seconddifference amplifiers as a function of a resistor connected betweenfirst and second outputs of the first and second difference amplifiers,respectively.
 18. The method of claim 15 wherein (a) comprises splittinga first current from a first output of the digital-to-analog converterand splitting a second current from a second output of thedigital-to-analog converter, and wherein (b) comprises driving theoutput amplifier with first and second differential amplifiers eachresponsive to split first and second currents.